Read the difference between mosfet and igbt

The internal structure of MOSFET and IGBT differs significantly, which directly influences their respective application fields. 1. Due to its structure, a MOSFET can typically handle large currents, even up to the kiloampere range, but its voltage blocking capability is not as strong as that of an IGBT. 2. On the other hand, IGBTs are capable of handling high power levels, with both high current and voltage capabilities. While their switching frequency isn't extremely high—usually around 100kHz—it's still quite good. However, compared to the MOSFET, which can operate at hundreds of kHz, MHz, or even tens of MHz, the IGBT's frequency range is more limited. This makes MOSFETs ideal for high-frequency applications like switching power supplies, ballasts, and high-frequency induction heating. 3. Based on their characteristics, MOSFETs are commonly used in high-frequency power supply applications such as switching power supplies, inverters, communication power supplies, and high-frequency welding machines. Meanwhile, IGBTs are mainly used in applications requiring high power, such as welding machines, inverters, electroplating systems, and high-power induction heating. The performance of Switch Mode Power Supplies (SMPS) heavily depends on the selection of power semiconductor devices, including switching transistors and rectifiers. Although there is no perfect solution for choosing between IGBT and MOSFET, comparing their performance in specific SMPS applications helps determine key parameters. This article discusses several important parameters, such as switching losses in hard-switching and soft-switching ZVS (Zero-Voltage Switching) topologies. It also describes three main power switching losses associated with circuit and device characteristics: conduction loss, turn-on loss, and turn-off loss. Additionally, it explores how the recovery characteristics of the diode play a critical role in determining the conduction switching loss of MOSFETs and IGBTs, and how this affects the performance of hard-switching topologies. **Conduction Loss** In addition to the longer voltage drop time of the IGBT, the turn-on characteristics of the IGBT and the power MOSFET are very similar. From the basic IGBT equivalent circuit (see Figure 1), it can be seen that the time required to fully adjust the minority carriers in the base region of the PNP BJT results in a voltage tail. This delay causes a quasi-saturation effect that prevents the collector/emitter voltage from dropping immediately to its VCE(sat) value. In ZVS cases, the VCE voltage may rise when the load current switches from the anti-parallel diode to the IGBT's collector. The Eon energy consumption listed in the IGBT product specification is the time integral of the product of the Icollector and VCE for each conversion cycle, measured in joules. It includes other losses related to saturation. It is further divided into two Eon energy parameters, Eon1 and Eon2. Eon1 refers to the power loss without considering the energy associated with the recovery of the hard-switched diode, while Eon2 includes the hard-switching conduction energy associated with diode recovery. The Eon2 test circuit is shown in Figure 2. The IGBT measures Eon by switching between two pulses. The first pulse increases the inductor current to the desired test current, and the second pulse measures the Eon loss during the diode recovery. In hard-switching circuits, the gate drive voltage, impedance, and the recovery characteristics of the rectifier diode determine the Eon switching loss. For a conventional CCM boost PFC circuit, the boost diode's recovery characteristics are crucial in controlling Eon energy consumption. Choosing a boost diode with minimal Trr and QRR, along with soft recovery characteristics, is essential. The tb/ta ratio has a significant impact on electrical noise and voltage spikes generated by the switching device. In hard-switching topologies like full-bridge and half-bridge circuits, the IGBT is often paired with a fast recovery transistor or a MOSFET body diode. When the corresponding switching transistor turns on, the diode carries current, and its recovery characteristics determine the Eon loss. Therefore, selecting a MOSFET with fast body diode recovery is important. However, the parasitic diode or body diode recovery characteristics of MOSFETs are generally slower than those of discrete diodes used in the industry. As a result, body diodes often limit the operating frequency of SMPS in hard-switching applications. IGBT packaged diodes are selected based on their application. Slower ultrafast diodes with lower forward conduction losses are often combined with low VCE(sat) motor drive IGBTs. Conversely, soft-recovery ultrafast diodes can be used with high-frequency SMPS IGBTs. In addition to selecting the right diode, designers can control Eon loss by adjusting the gate drive source impedance. Reducing the drive source impedance increases the conduction di/dt and reduces Eon loss. However, there is a trade-off between Eon loss and EMI, as higher di/dt leads to increased voltage spikes and EMI. To find the correct gate drive impedance, internal circuit testing and verification may be necessary, and approximate values can be determined from the MOSFET transfer curve. Assuming the FET current rises to 10A when turned on, according to the 25°C curve in Figure 3, the gate voltage must increase from 5.2V to 6.7V to reach 10A. The average GFS is calculated as 10A / (6.7V - 5.2V) = 6.7mΩ. Using this average GFS value, the gate drive impedance required to achieve a specific di/dt can be calculated. Applying the average GFS value to the equation, we get the gate drive impedance. Similarly, IGBTs can also undergo similar gate drive resistance calculations. VGE(avg) and GFS can be determined from the IGBT's transfer characteristic curve, and the CIES value under VGE(avg) is used instead of Ciss. The calculated IGBT turn-on gate drive impedance is 100Ω, which is higher than the previous 37Ω, indicating that the IGBT has a higher GFS and lower CIES. The key point here is that the gate drive circuit must be adjusted when switching from MOSFET to IGBT. **Conduction losses need to be cautious** When comparing devices rated at 600V, the conduction losses of IGBTs are generally lower than those of 600V MOSFETs of the same chip size. This comparison should be made at the collector and drain current densities and performed at the worst-case operating junction temperature. For example, both the FGP20N6S2 SMPS2 IGBT and the FCP11N60 SuperFET have an RθJC value of 1°C/W. Figure 4 shows the conduction loss versus DC current at a junction temperature of 125°C. The plot indicates that the MOSFET has higher conduction loss after the DC current exceeds 2.92A. However, the DC conduction losses in Figure 4 are not suitable for most applications. A comparison curve of conduction loss in CCM (continuous current mode), boost PFC circuit, junction temperature of 125°C, and AC input voltage Vac and 400Vdc DC output voltage of 85V is shown in Figure 5. In the figure, the intersection point of the MOSFET-IGBT curve is 2.65A RMS. For PFC circuits, the MOSFET has a large conduction loss when the AC input current is greater than 2.65A RMS. The 2.65A PFC AC input current corresponds to 2.29A RMS calculated using Equation 2 in the MOSFET. The MOSFET conduction loss, I²R, can be calculated using the current defined by Equation 2 and the RDS(on) of the MOSFET at 125°C. Taking into account the variation of RDS(on) with the drain current, the conduction loss can be further refined. This relationship is shown in Figure 6. An IEEE article titled "How to Incorporate the RDS(on) of a Power MOSFET to the Drain Current Transient Value into the Conduction Loss Calculation of a High-Frequency Three-Phase PWM Inverter" describes how to determine the effect of drain current on conduction losses. As a function of ID, changes in RDS(on) have little effect on most SMPS topologies. For example, in a PFC circuit, when the peak current ID of the FCP11N60 MOSFET is 11A—doubled to 5.5A (the test condition for RDS(on) in the specification), the effective value and conduction loss of RDS(on) increase by 5%. In high-pulse-current topologies where the MOSFET conducts for very small duty cycles, the characteristics shown in Figure 6 should be considered. If the FCP11N60 MOSFET is operating in a circuit with a 20A pulse and a 7.5% duty cycle (i.e., 5.5A RMS), then the effective RDS(on) will be greater than 5.5A (test current in the specification). 0.32 ohms is 25% larger. Equation 2: RMS current in CCM PFC circuit In Equation 2, Iacrms is the RMS input current of the PFC circuit; Vac is the RMS input voltage of the PFC circuit; Vout is the DC output voltage. In practical applications, calculating the conduction losses of IGBTs in similar PFC circuits is more complicated because each switching cycle involves different ICs. The VCE(sat) of an IGBT cannot be represented by an impedance. A relatively straightforward method is to represent it as an impedance RFCE in series with a fixed VFCE voltage, VCE(ICE) = ICE × RFCE + VFCE. Thus, the conduction loss can be calculated as the product of the average collector current and VFCE, plus the square of the RMS collector current multiplied by the impedance RFCE. The example in Figure 5 only considers the conduction losses of the CCM PFC circuit, assuming that the design goal is to maintain a worst-case conduction loss of less than 15W. Taking the FCP11N60 MOSFET as an example, the circuit is limited to 5.8A, while the FGP20N6S2 IGBT can operate at 9.8A AC input current. It can conduct more than 70% of the power of the MOSFET. Although the conduction loss of IGBTs is small, most 600V IGBTs are PT (Punch Through) type devices. PT devices have NTC (negative temperature coefficient) characteristics and cannot be shunted in parallel. Perhaps these devices can be paralleled with limited effectiveness through matching devices VCE(sat), VGE(TH) (gate threshold voltage), and mechanical packaging to allow consistent temperature variations in IGBT chips. Conversely, MOSFETs have a PTC (Positive Temperature Coefficient) that provides good current shunting. **Shutdown Loss** In hard-switching, clamp-inductive circuits, the turn-off loss of the MOSFET is much lower than that of the IGBT due to the tail current of the IGBT, which is related to the removal of the minority carriers of the PNP BJT in Figure 1. Figure 7 shows a function Eoff of collector current ICE and junction temperature Tj, the curve of which is provided in most IGBT data sheets. These curves are based on clamped inductive circuits and have the same test voltage and contain tail current energy losses. Figure 2 shows a typical test circuit for measuring IGBT Eoff. Its test voltage, VDD in Figure 2, varies from manufacturer to manufacturer and from individual device BVCES. VDD in this test condition should be considered when comparing devices, as testing and operation at lower VDD clamp voltages will result in reduced Eoff power consumption. Reducing the gate drive turn-off impedance has minimal effect on reducing the IGBT Eoff loss. As shown in FIG. 1, when the equivalent majority carrier MOSFET is turned off, there is still a storage time delay td(off) I in the IGBT minority carrier BJT. However, lowering the Eoff drive impedance will reduce the risk of the current caused by the Miller capacitance CRES and the dv/dt turning off the VCE into the gate drive loop, avoiding re-biasing the device to a conducting state, resulting in multiple switching actions that produce Eoff. The ZVS and ZCS topologies have advantages in reducing the turn-off losses of MOSFETs and IGBTs. However, the working advantage of ZVS is not so large in IGBTs, because the tailing surge current Eoff is caused when the collector voltage rises to a potential value that allows excess stored charge to be dissipated. The ZCS topology boosts maximum IGBT Eoff performance. The correct gate drive sequence allows the IGBT gate signal to not be cleared before the second collector current zero crossing, thereby significantly reducing the IGBT ZCS Eoff. The Eoff energy dissipation of a MOSFET is a function of its Miller capacitance Crss, gate drive speed, gate drive turn-off source impedance, and parasitic inductance in the source power circuit path. The circuit's parasitic inductance Lx (shown in Figure 8) produces an electrical potential that increases the turn-off loss by limiting the current speed drop. At turn-off, the current drop rate di/dt is determined by Lx and VGS(th). If Lx = 5nH and VGS(th) = 4V, the maximum current falling speed is VGS(th) / Lx = 800A/μs. **To Sum Up** When choosing a power switching device, there is no one-size-fits-all solution. Circuit topology, operating frequency, ambient temperature, and physical size all play a role in making the best choice. In ZVS and ZCS applications with minimal Eon loss, MOSFETs can operate at higher frequencies due to faster switching speeds and less turn-off losses. For hard-switching applications, the recovery characteristics of MOSFET parasitic diodes can be a disadvantage. In contrast, since the diodes in the IGBT package are matched to the specific application, an excellent soft recovery diode can be matched to the higher-speed SMPS device. Afterword: There is no essential difference between MOSFET and IGBT. The question of whether MOSFET is better or IGBT is often incorrect. As for why we sometimes use MOSFETs and sometimes not, we can't simply distinguish between good and bad. To judge, we need to use a dialectical approach to consider this issue.

FPV Drone Frame

3K Carbon Fiber drone Frame,Quadcopter Freestyle RC FPV Drone Frame,FPV Drone Frame 7inch

Jiangsu Yunbo Intelligent Technology Co., Ltd , https://www.fmodel-ai.com